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ISL88731A
Data Sheet January 7, 2009 FN6738.1
SMBus Level 2 Battery Charger
The ISL88731A is a highly integrated Lithium-ion battery charger controller, programmable over the SMBus system management bus (SMBus). The ISL88731A is intended to be used in a smart battery charger (SBC) within a smart battery system (SBS) that throttles the charge power such that the current from the AC-adapter is automatically limited. High efficiency is achieved with a DC/DC synchronous-rectifier buck converter, equipped with diode emulation for enhanced light load efficiency and system bus boosting prevention. The ISL88731A charges one to four Lithium-ion series cells, and delivers up to 8A charge current. Integrated MOSFET drivers and bootstrap diode result in fewer components and smaller implementation area. Low offset current-sense amplifiers provide high accuracy with 10m sense resistors. The ISL88731A provides 0.5% end-of-charge battery voltage accuracy. The ISL88731A provides a digital output that indicates the presence of the AC-adapter as well as an analog output which indicates the adapter current within 4% accuracy. The ISL88731A is available in a small 5mmx5mm 28 Ld thin (0.8mm) QFN package. An evaluation kit is available to reduce design time. The ISL88731A is available in Pb-Free packages.
Features
* 0.5% Battery Voltage Accuracy * 3% Adapter Current Limit Accuracy * 3% Charge Current Accuracy * SMBus 2 Wire Serial Interface * Battery Short Circuit Protection * Fast Response for Pulse-Charging * Fast System-Load Transient Response * Monitor Outputs - Adapter Current (3% Accuracy) - AC-Adapter Detection * 11-Bit Battery Voltage Setting * 6 Bit Charge Current/Adapter Current Setting * 8A Maximum Battery Charger Current * 11A Maximum Adapter Current * +8V to +28V Adapter Voltage Range * Pb-Free (RoHS compliant)
Applications
* Notebook Computers * Tablet PCs
Pinout
ISL88731A (28 LD TQFN) TOP VIEW
PHASE UGATE BOOT CSSN CSSP DCIN VCC
* Portable Equipment with Rechargeable Batteries
Ordering Information
PART NUMBER (Note)
21 20 19 18 17 16 15 VDDP
28 NC ACIN VREF ICOMP NC VCOMP NC 1 2 3 4 5 6 7 8 ICM
27
26
25
24
23
22
PART MARKING ISL887 31AHRZ
TEMP RANGE (C)
PACKAGE (Pb-Free)
PKG. DWG. # L28.5x5B L28.5x5B
ISL88731AHRZ
LGATE PGND CSOP CSON NC VFB
-10 to +100 28 Ld 5x5 TQFN -10 to +100 28 Ld 5x5 TQFN (Tape and Reel)
ISL88731AHRZ-T* ISL887 31AHRZ
9 SDA
10 SCL
11 VDDSMB
12 GND
13 ACOK
14 NC
*Please refer to TB347 for details on reel specifications. NOTE: These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright Intersil Americas Inc. 2008, 2009. All Rights Reserved All other trademarks mentioned are the property of their respective owners.
ISL88731A
DCIN
VCC
VDDSMB SDA SCL SMBUS
11 6 6
DACV DACS DACI EN
DACV DACS DACI
LINEAR REGULATOR
VDDP
REFERENCE
VREF
REF
+ -
ACOK
ACIN
BUFF
ICM EN BOOT
CSSP CSSN ICOMP
LEVEL SHIFTER 20x
DACS GMS
+
CSO UGATE
CSOP CSON
LEVEL SHIFTER 20x
GMI DACI
+
LVB DACV
DC/DC CONVERTER
PHASE VDDP LGATE PGND
+
GMV
LVB
VFB 500k VCOMP 100k EN CSSP
GND
FIGURE 1. FUNCTIONAL BLOCK DIAGRAM
AC ADAPTER
RS1
TO SYSTEM
CSSP ACIN DCIN
CSSN UGATE PHASE RS2 TO BATTERY
ISL88731A
BOOT LGATE AGND ICOMP VCOMP VDDP VREF CSOP CSON VFB PGND
VCC GND
ACOK ICM SDA SCL VDDSMB
PGND HOST
AGND
FIGURE 2. TYPICAL APPLICATION CIRCUIT
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FN6738.1 January 7, 2009
ISL88731A
Absolute Maximum Ratings
DCIN, CSSP, CSSN, CSOP, CSON, VFB . . . . . . . . . . -0.3V to +28V CSSP-CSSN, CSOP-CSON, PGND-GND. . . . . . . . . -0.3V to +0.3V PHASE to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -6V to +30V BOOT to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V BOOT to PHASE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6V UGATE . . . . . . . . . . . . . . . . . . . . . . .PHASE - 0.3V to BOOT +0.3V LGATE . . . . . . . . . . . . . . . . . . . . . . . .PGND - 0.3V to VDDP +0.3V ICOMP, VCOMP, VREF, to GND . . . . . . . . . . . . -0.3V to VCC +0.3V VDDSMB, SCL, SDA, ACIN, ACOK . . . . . . . . . . . . . . . -0.3V to +6V VDDP, ICM, VCC to GND, VDDP to PGND. . . . . . . . . . -0.3V to +6V
Thermal Information
Thermal Resistance (Typical, Notes 1, 2) JA (C/W) JC (C/W) QFN Package. . . . . . . . . . . . . . . . . . . . 39 9.5 Junction Temperature Range. . . . . . . . . . . . . . . . . .-55C to +150C Operating Temperature Range . . . . . . . . . . . . . . . .-10C to +100C Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . .-65C to +150C Pb-Free Reflow Profile. . . . . . . . . . . . . . . . . . . . . . . . .see link below http://www.intersil.com/pbfree/Pb-FreeReflow.asp
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty.
NOTES: 1. JA is measured with the component mounted on a highly effective thermal conductivity test board on free air. See Tech Brief TB379 for details. 2. For JC, the "case temp" location is the center of the exposed metal pad on the package underside.
Electrical Specifications
DCIN = CSSP = CSSN = 18V, CSOP = CSON = 12V, VDDP = 5V, BOOT-PHASE = 5.0V, GND = PGND = 0V, CVDDP = 1F, IVDDP = 0mA, TA = -10C to +100C. Parameters with MIN and/or MAX limits are 100% tested at +25C, unless otherwise specified. Temperature limits established by characterization and are not production tested. CONDITIONS MIN TYP MAX UNITS
PARAMETER CHARGE VOLTAGE REGULATION
Battery Full Charge Voltage and Accuracy ChargeVoltage = 0x41A0
16.716 -0.5
16.8
16.884 0.5
V % V % V % V % V mV
ChargeVoltage = 0x3130
12.529 -0.5
12.592
12.655 0.5
ChargeVoltage = 0x20D0
8.350 -0.6
8.4
8.450 0.6
ChargeVoltage = 0x1060
4.163 -0.7
4.192
4.221 0.7
Battery Undervoltage Lockout Trip Point for Trickle Charge Battery Undervoltage Lockout Trip Point Hysteresis CHARGE CURRENT REGULATION CSOP to CSON Full-Scale Current-Sense Voltage Charge Current and Accuracy
VFB rising
2.55 100
2.7 250
2.85 400
78.22 RS2 = 10m (see Figure 2) ChargingCurrent = 0x1f80 RS2 = 10m (see Figure 2) ChargingCurrent = 0x0f80 RS2 = 10m (see Figure 2) ChargingCurrent = 0x0080 7.822 -3 3.809 -4 64 -1.6 0
80.64 8.064
83.06 8.306 3
mV A % A % mA % V
3.968
4.126 4
128
220 1.4 19
Charge Current Gain Error CSOP/CSON Input Voltage Range
Based on charge current = 128mA and 8.064A
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FN6738.1 January 7, 2009
ISL88731A
Electrical Specifications
DCIN = CSSP = CSSN = 18V, CSOP = CSON = 12V, VDDP = 5V, BOOT-PHASE = 5.0V, GND = PGND = 0V, CVDDP = 1F, IVDDP = 0mA, TA = -10C to +100C. Parameters with MIN and/or MAX limits are 100% tested at +25C, unless otherwise specified. Temperature limits established by characterization and are not production tested. (Continued) CONDITIONS Adapter present, not charging, ICSOP + ICSON + IPHASE + ICSSP + ICSSN + IFB VPHASE = VCSON = VCSOP = VDCIN = 19V, VACIN = 5V Adapter Absent ICSOP + ICSON + IPHASE + ICSSP + ICSSN + IFB VPHASE = VCSON = VCSOP = 19V, VDCIN = 0V Adapter Quiescent Current INPUT CURRENT REGULATION CSSP to CSSN Full-Scale Current-Sense CSSP = 19V Voltage Input Current Accuracy RS1 = 10m (see Figure 2) Adapter Current = 11004mA or 3584mA RS1 = 10m (see Figure 2) Adapter Current = 2048mA Input Current Limit Gain Error Input Current Limit Offset CSSP/CSSN Input Voltage Range ICM Gain ICM Offset ICM Accuracy VCSSP-CSSN = 110mV Based on VCSSP-CSSN = 110mV and 20mV VCSSP-CSSN = 110mV VCSSP-CSSN = 55mV or 35mV VCSSP-CSSN = 20mV ICM Max Output Current SUPPLY AND LINEAR REGULATOR DCIN, Input Voltage Range VDDP Output Voltage VDDP Load Regulation VDDSMB Range VDDSMB UVLO Rising VDDSMB UVLO Hysteresis VDDSMB Quiescent Current V REFERENCE VREF Output Voltage ACOK ACOK Sink Current ACOK Leakage Current ACIN ACIN rising Threshold ACIN Threshold Hysteresis ACIN Input Bias Current 3.1 40 -1 3.2 60 3.3 90 1 V mV A VACOK = 0.4V, ACIN = 1.5V VACOK = 5.5V, ACIN = 2.5V 2 8 1 mA A 0 < IVREF < 300A 3.1 3.2 3.3 V VDDP = SCL = SDA = 5.5V 8.0V < VDCIN < 28V, no load 0 < IVDDP < 30mA 2.7 2.4 100 2.5 150 20 8 4.9 5.1 35 26 5.3 100 5.5 2.6 200 27 V V mV V V mV A VCSSP-CSSN = 0.1V Based on InputCurrent = 1024mA and 11004mA 106.7 -3 -5 -1.5 -1 8 19.8 -1.5 -2.5 -4 -8 20 110 113.3 3 5 1.5 1 26 20.2 1.5 2.5 4 8 500 mV % % % mV V V/V mV % % % A IDCIN+ICSSP+ICSSN VADAPTER = 8V to 26V, VBATTERY 4V to 16.8V -1 MIN TYP 135 MAX 400 UNITS A
PARAMETER Battery Quiescent Current
0.2
4
A
3
10
mA
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FN6738.1 January 7, 2009
ISL88731A
Electrical Specifications
DCIN = CSSP = CSSN = 18V, CSOP = CSON = 12V, VDDP = 5V, BOOT-PHASE = 5.0V, GND = PGND = 0V, CVDDP = 1F, IVDDP = 0mA, TA = -10C to +100C. Parameters with MIN and/or MAX limits are 100% tested at +25C, unless otherwise specified. Temperature limits established by characterization and are not production tested. (Continued) CONDITIONS MIN TYP MAX UNITS
PARAMETER SWITCHING REGULATOR Frequency BOOT Supply Current PHASE Input Bias Current UGATE ON-Resistance Low UGATE ON-Resistance High LGATE ON-Resistance High LGATE ON-Resistance Low Dead Time ERROR AMPLIFIERS GMV Amplifier Transconductance GMI Amplifier Transconductance GMS Amplifier Transconductance GMI/GMS Saturation Current GMV Saturation Current ICOMP, VCOMP Clamp Voltage LOGIC LEVELS SDA/SCL Input Low Voltage SDA/SCL Input High Voltage SDA/SCL Input Bias Current SDA, Output Sink Current UGATE High
330 170
400 290 0 0.9 1.4 1.4 0.9
440 400 2 1.6 2.5 2.5 1.6 80
kHz A A ns
VDCON = 28V, VCSON = VPHASE = 20V IUGATE = -100mA (Note 4) IUGATE = 10mA (Note 4) ILGATE = +10mA (Note 4) ILGATE = -100mA (Note 4) Falling UGATE to rising LGATE or falling LGATE to rising UGATE 35
50
200 40 40 15 10 0.25V < VICOMP, VCOMP < 3.5V 200
250 50 50 21 17 300
300 60 60 25 30 400
A/V A/V A/V A A mV
VDDSMB = 2.7V to 5.5V VDDSMB = 2.7V to 5.5V VDDSMB = 2.7V to 5.5V VSDA = 0.4V 2 -1 7 15
0.8
V V
1
A mA
SMBus Timing Specification VDDSMB = 2.7V TO 5.5V
PARAMETERS SMBus Frequency Bus Free Time Start Condition Hold Time from SCL Start Condition Setup Time from SCL Stop Condition Setup Time from SCL SDA Hold Time from SCL SDA Setup Time from SCL SCL Low Timeout (Note 3) SCL Low Period SCL High Period Maximum Charging Period Without a SMBus Write to ChargeVoltage or ChargeCurrent Register NOTES: 3. If SCL is low for longer than the specified time, the charger is disabled. 4. Limits established by characterization and are not production tested. SYMBOL FSMB TBUF THD:STA TSU:STA TSU:STO THD:DAT TSU:DAT TTIMEOUT TLOW THIGH CONDITIONS MIN 10 4.7 4 4.7 4 300 250 22 4.7 4 140 180 220 25 30 TYP MAX 100 UNITS kHz s s s s ns ns ms s s s
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FN6738.1 January 7, 2009
ISL88731A Typical Operating Performance
5.15
DCIN = 20V, 3S2P Li-Battery, TA = +25C, unless otherwise noted.
3.23 3.22 0.5% 1.0%
5.10
5.05 VDDP (V) VREF (V)
3.21
5.00
3.20 3.19
0.0%
4.95
-0.5% 4.90 4.85 0 20 40 60 80 100 VDDP LOAD CURRENT (mA) 3.18 3.17 0 50 100 I VREF (A) 150 -1.0% 200
FIGURE 3. VDD LOAD REGULATION
FIGURE 4. VREF LOAD REGULATION
15
13.0
3.5 3.0 2.5 BATTERY CURRENT
10 ICM ACCURACY (%) BATTERY VOLTAGE
12.5
5
12.0
VCHG (V) 2.0
0
11.5 1.5 11.0 1.0 10.5 ICHG (A) 0.5 0.0 160
-5
-10
-15
1
2
3
4
5
6
7
8
10.0
0
20
40
ADAPTER CURRENT (A)
60 80 100 TIME (MINUTES)
120
140
FIGURE 5. ICM ACCURACY vs AC-ADAPTER CURRENT
FIGURE 6. TYPICAL CHARGING VOLTAGE AND CURRENT
VCOMP ICOMP
ICOMP VCOMP
CHARGE CURRENT CHARGE CURRENT INDUCTOR CURRENT INDUCTOR CURRENT
FIGURE 7. CHARGE ENABLE
FIGURE 8. CHARGE DISABLE
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FN6738.1 January 7, 2009
ISL88731A Typical Operating Performance
UGATE
DCIN = 20V, 3S2P Li-Battery, TA = +25C, unless otherwise noted. (Continued)
LGATE
UGATE
INDUCTOR CURRENT
PHASE INDUCTOR CURRENT
LGATE
PHASE
FIGURE 9. SWITCHING WAVEFORMS AT DIODE EMULATION
FIGURE 10. SWITCHING WAVEFORMS IN CC MODE
CSON/ VBATTERY CSON/ VBATTERY
BATTERY CURRENT
BATTERY CURRENT
FIGURE 11. BATTERY REMOVAL
FIGURE 12. BATTERY INSERTION
100
SYSTEM LOAD BATTERY VOLTAGE CHARGE CURRENT
95
90 ()
16.8VBATTERY
85 12.6VBATTERY 80 ADAPTER CURRENT 8.4VBATTERY
75 70 0
4.2VBATTERY 2 4 CHARGE CURRENT (A) 6 8
FIGURE 13. LOAD TRANSIENT RESPONSE
FIGURE 14. EFFICIENCY vs CHARGE CURRENT AND BATTERY VOLTAGE (EFFICIENCY DCIN = 20V)
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FN6738.1 January 7, 2009
ISL88731A Functional Pin Descriptions
BOOT
High-Side Power MOSFET Driver Power-Supply Connection. Connect a 0.1F capacitor from BOOT to PHASE.
PGND
Power Ground. Connect PGND to the source of the low side MOSFET.
VCC
Power input for internal analog circuits. Connect a 4.7 resistor from VCC to VDDP and a 1F ceramic capacitor from VCC to ground.
UGATE
High-Side Power MOSFET Driver Output. Connect to the high-side N-Channel MOSFET gate.
VDDP
Linear Regulator Output. VDDP is the output of the 5.2V linear regulator supplied from DCIN. VDDP also directly supplies the LGATE driver and the BOOT strap diode. Bypass with a 1F ceramic capacitor from VDDP to PGND.
LGATE
Low-Side Power MOSFET Driver Output. Connect to low-side N-Channel MOSFET. LGATE drives between VDDP and PGND.
PHASE
High-Side Power MOSFET Driver Source Connection. Connect to the source of the high-side N-Channel MOSFET.
ICOMP
Compensation Point for the charging current and adapter current regulation Loop. Connect 0.01F to GND. See "Voltage Control Loop" on page 20 for details of selecting the ICOMP capacitor.
CSOP
Charge Current-Sense Positive Input.
VCOMP
Compensation Point for the voltage regulation loop. Connect 4.7k in series with 0.01F to GND. See "Voltage Control Loop" on page 20 for details on selecting VCOMP components.
CSON
Charge Current-Sense Negative Input.
CSSP
Input Current-Sense Positive Input.
VFB
Feedback for the Battery Voltage.
CSSN
Input Current-Sense Negative Input.
VDDSMB
SMBus interface Supply Voltage Input. Bypass with a 0.1F capacitor to GND.
DCIN
Charger Bias Supply Input. Bypass DCIN with a 0.1F capacitor to GND.
SDA
SMBus Data I/O. Open-drain Output. Connect an external pull-up resistor according to SMBus specifications.
ACIN
AC-adapter Detection Input. Connect to a resistor divider from the AC-adapter output.
SCL
SMBus Clock Input. Connect an external pull-up resistor according to SMBus specifications.
ACOK
AC Detect Output. This open drain output is high impedance when ACIN is greater than 3.2V. The ACOK output remains low when the ISL88731A is powered down. Connect a 10k pull-up resistor from ACOK to VDDSMB.
GND
Analog Ground. Connect directly to the backside paddle. Connect to PGND close to the output capacitor.
ICM
Input Current Monitor Output. ICM voltage equals 20 x (VCSSP - VCSSN).
Back Side Paddle
Connect the backside paddle to GND.
NC
No Connect. Pins 1, 5, 7 and 14 are not connected.
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FN6738.1 January 7, 2009
ISL88731A Theory of Operation
Introduction
The ISL88731A includes all of the functions necessary to charge 1-to-4 cell Li-ion and Li-polymer batteries. A high efficiency synchronous buck converter is used to control the charging voltage up to 19.2V and charging current up to 8A. The ISL88731A also has input current limiting up to 11A. The Input current limit, charge current limit and charge voltage limit are set by internal registers written with SMBus. Figure 2 shows the ISL88731A "Typical Application Circuit". The ISL88731A charges the battery with constant charge current, set by the ChargeCurrent register, until the battery voltage rises to a voltage set by the ChargeVoltage register. The charger will then operate at a constant voltage. The adapter current is monitored and if the adapter current rises to the limit set by the InputCurrent register, battery charge current is reduced so the charger does not reduce the adapter current available to the system. The ISL88731A features a voltage regulation loop (VCOMP) and two current regulation loops (ICOMP). The VCOMP voltage regulation loop monitors VFB to limit the battery charge voltage. The ICOMP current regulation loop limits the battery charging current delivered to the battery to ensure that it never exceeds the current set by the ChargeCurrent register. The ICOMP current regulation loop also limits the input current drawn from the AC-adapter to ensure that it never exceeds the limit set by the InputCurrent register, and to prevent a system crash and AC-adapter overload. An adaptive gate drive scheme is used to control the dead time between two switches. The dead time control circuit monitors the LGATE output and prevents the upper side MOSFET from turning on until 20ns after LGATE falls below 1V VGS, preventing cross-conduction and shoot-through. The same occurs for LGATE turn on. In order for the dead time circuit to work properly, there must be a low resistance, low inductance path from the LGATE driver to MOSFET gate, and from the source of MOSFET to PGND. An internal Schottky diode between the VDDP pin and BOOT pin keeps the bootstrap capacitor charged.
AC-Adapter Detection
Connect the AC-adapter voltage through a resistor divider to ACIN to detect when AC power is available, as shown in Figure 2. ACOK is an open-drain output and is active low when ACIN is less than Vth,fall, and high when ACIN is above Vth,rise. The ACIN rising threshold is 3.2V (typ) with 57mV hysteresis.
Current Measurement
Use ICM to monitor the adapter current being sensed across CSSP and CSSN. The output voltage range is 0V to 2.5V. The voltage of ICM is proportional to the voltage drop across CSSP and CSSN, and is given by Equation 1:
ICM = 20 I INPUT R S1 (EQ. 1)
where IADAPTER is the DC current drawn from the AC-adapter. It is recommended to have an RC filter at the ICM output for minimizing the switching noise.
PWM Control
The ISL88731A employs a fixed frequency PWM control architecture with a feed-forward function. The feed-forward function maintains a constant modulator gain of 11 to achieve fast line regulation as the input voltage changes. The duty cycle of the buck regulator is controlled by the lower of the voltages on ICOMP and VCOMP. The voltage on ICOMP and VCOMP are inputs to a Lower Voltage Buffer (LVB) who's output is the lower of the two inputs. The output of the LVB is compared to an internal 400kHz ramp to produce the Pulse Width Modulated signal that controls the UGATE and LGATE drivers. An internal clamp holds the higher of the two voltages (0.3V) above the lower voltage. This speeds the transition from voltage loop control to current loop control or vice versa. The ISL88731A can operate up to 99.6% duty cycle if the input voltage drops close to or below the battery charge voltage (drop out mode). The DC/DC converter has a timer to prevent the frequency from dropping into the audible frequency range. To prevent boosting of the system bus voltage, the battery charger drives the lower FET in a way that prevents negative inductor current.
VDDP Regulator
VDDP provides a 5.2V supply voltage from the internal LDO regulator from DCIN and can deliver up to 30mA of continuous current. The MOSFET drivers are powered by VDDP. VDDP also supplies power to VCC through a low-pass filter as shown in the "Typical Application Circuit" (see Figure 2) on page 2. Bypass VDDP and VCC with a 1F capacitor.
VDDSMB Supply
The VDDSMB input provides power to the SMBus interface. Connect VDDSMB to VCC, or apply an external supply to VDDSMB to keep the SMBus interface active while the supply to DCIN is removed. When VDDSMB is biased the internal registers are maintained. Bypass VDDSMB to GND with a 0.1F or greater ceramic capacitor.
Short Circuit Protection and 0V Battery Charging
Since the battery charger will regulate the charge current to the limit set by the ChargeCurrent register, it automatically has short circuit protection and is able to provide the charge current to wake up an extremely discharged battery. Undervoltage trickle charge folds back current if there is a short circuit on the output.
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FN6738.1 January 7, 2009
ISL88731A
Undervoltage Detect and Battery Trickle Charging
If the voltage at CSON falls below 2.5V, the ISL88731A reduces the charge current limit to 128mA to trickle charge the battery. When the voltage rises above 2.7V the charge current reverts to the programmed value in the ChargeCurrent register.
Overvoltage Protection
ISL88731 has an Overvoltage Protection circuit that limits the output voltage when the battery is removed or disconnected by a pulse charging circuit. If CSON exceeds the output voltage set point in the chargevoltage register by more than 300mV an internal comparator pulls VCOMP down and turns off both upper and lower FETs of the buck as in Figure 15. There is a delay of approximately 1s between VOUT exceeding the OVP trip point and pulling VCOMP, LGATE and UGATE low. After UGATE and LGATE are turned OFF inductor current continues to flow through the body diode of the lower FET and VOUT continues to rise until inductor current reaches zero.
Over-Temperature Protection
If the die temperature exceeds +150C, it stops charging. Once the die temperature drops below +125C, charging will start up again.
FIGURE 15. OVERVOLTAGE PROTECTION IN ISL88731
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FN6738.1 January 7, 2009
ISL88731A
The System Management Bus
The System Management Bus (SMBus) is a 2-wire bus that supports bidirectional communications. The protocol is described briefly here. More detail is available from www.smbus.org.
START and STOP Conditions
As shown in Figure 17, START condition is a HIGH-to-LOW transition of the SDA line while SCL is HIGH. The STOP condition is a LOW-to-HIGH transition on the SDA line while SCL is HIGH. A STOP condition must be sent before each START condition.
General SMBus Architecture
VDDSMB SMBUS SLAVE SCL OUTPUTCONTROL INPUT SDA OUTPUTCONTROL INPUT STATE MACHINE, REGISTERS, MEMORY, ETC
SDA
SMBUS MASTER INPUT SCL CONTROL OUTPUT CPU SDA CONTROL OUTPUT INPUT
SCL S P STOP CONDITION START CONDITION
SMBUS SLAVE SCL OUTPUT CONTROL S CL SDA OUTPUT CONTROL INPUT INPUT STATE MACHINE, REGISTERS, MEMORY , ETC
FIGURE 17. START AND STOP WAVEFORMS
Acknowledge
Each address and data transmission uses 9-clock pulses. The ninth pulse is the acknowledge bit (ACK). After the start condition, the master sends 7-slave address bits and a R/W bit during the next 8-clock pulses. During the ninth clock pulse, the device that recognizes its own address holds the data line low to acknowledge. The acknowledge bit is also used by both the master and the slave to acknowledge receipt of register addresses and data (see Figure 18).
SCL 1 SDA MSB 2 8 9
TO OTHER SLAVE DEVICES
Data Validity
The data on the SDA line must be stable during the HIGH period of the SCL, unless generating a START or STOP condition. The HIGH or LOW state of the data line can only change when the clock signal on the SCL line is LOW. Refer to Figure 16.
SDA
SDA
START
ACKNOWLEDGE FROM SLAVE
SCL DATA LINE CHANGE STABLE OF DATA DATA VALID ALLOWED
FIGURE 18. ACKNOWLEDGE ON THE I2C BUS
SMBus Transactions
All transactions start with a control byte sent from the SMBus master device. The control byte begins with a Start condition, followed by 7-bits of slave address (0001001 for the ISL88731A) followed by the R/W bit. The R/W bit is 0 for a write or 1 for a read. If any slave devices on the SMBus bus recognize their address, they will Acknowledge by pulling the serial data (SDA) line low for the last clock cycle in the control byte. If no slaves exist at that address or are not ready to communicate, the data line will be 1, indicating a Not Acknowledge condition. Once the control byte is sent, and the ISL88731A acknowledges it, the 2nd byte sent by the master must be a register address byte such as 0x14 for the ChargeCurrent register. The register address byte tells the ISL88731A which register the master will write or read. See Table 1 for details of the registers. Once the ISL88731A receives a register address byte it responds with an acknowledge.
FIGURE 16. DATA VALIDITY
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FN6738.1 January 7, 2009
ISL88731A
Write To A Register
S SLAVE ADDR + W A REGISTER ADDR A LO BYTE DATA A HI BYTE DATA A P
Read From A Register
S SLAVE ADDR + W S P START STOP A REGISTER ADDR A N A P S SLAVE ADDR + R A LO BYTE DATA A HI BYTE DATA N P
ACKNOWLEDGE NO ACKNOWLEDGE
DRIVEN BY THE MASTER DRIVEN BY ISL88731
FIGURE 19. SMBus/ISL88731A READ AND WRITE PROTOCOL
Byte Format
Every byte put on the SDA line must be eight bits long and must be followed by an acknowledge bit. Data is transferred with the most significant bit first (MSB) and the least significant bit last (LSB).
The data (SDA) and clock (SCL) pins have Schmitt-trigger inputs that can accommodate slow edges. Choose pull-up resistors for SDA and SCL to achieve rise times according to the SMBus specifications. The ISL88731A is controlled by the data written to the registers described in Table 1.
ISL88731A and SMBus
The ISL88731A receives control inputs from the SMBus interface. The serial interface complies with the SMBus protocols as documented in the System Management Bus Specification V1.1, which can be downloaded from www.smbus.org. The ISL88731A uses the SMBus Read-Word and Write-Word protocols (Figure 19) to communicate with the smart battery. The ISL88731A is an SMBus slave device and does not initiate communication on the bus. It responds to the 7-bit address 0b0001001_ (0x12). Read address = 0b00010011 and Write address = 0b00010010. In addition, the ISL88731A has two identification (ID) registers: a 16-bit device ID register and a 16-bit manufacturer ID register.
Battery Charger Registers
The ISL88731A supports five battery-charger registers that use either Write-Word or Read-Word protocols, as summarized in Table 1. ManufacturerID and DeviceID are "read only" registers and can be used to identify the ISL88731A. On the ISL88731A, ManufacturerID always returns 0x0049 (ASCII code for "I" for Intersil) and DeviceID always returns 0x0001.
Enabling and Disabling Charging
After applying power to ISL88731A, the internal registers contain their POR values (see Table 1). The POR values for charge current and charge voltage are 0x0000. These values disable charging. To enable charging, the ChargeCurrent register must be written with a number >0x007F and the ChargeVoltage register must be written with a number >0x000F. Charging can be disabled by writing 0x0000 to either of these registers.
TABLE 1. BATTERY CHARGER REGISTER SUMMARY REGISTER ADDRESS
0x14 0x15 0x3F 0xFE 0xFF
REGISTER NAME
ChargeCurrent ChargeVoltage InputCurrent ManufacturerID DeviceID
READ/WRITE
Read or Write Read or Write Read or Write Read Only Read Only
DESCRIPTION
6-bit Charge Current Setting 11-bit Charge Voltage Setting 6-bit Charge Current Setting Manufacturer ID Device ID
POR STATE
0x0000 0x0000 0x0080 0x0049 0x0001
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ISL88731A Setting Charge Voltage
Charge voltage is set by writing a valid 16-bit number to the ChargeVoltage register. This 16-bit number translates to a 65.535V full-scale voltage. The ISL88731A ignores the first 4 LSBs and uses the next 11 bits to set the voltage DAC. The charge voltage range of the ISL88731A is 1.024V to 19.200V. Numbers requesting charge voltage greater than 19.200V result in a ChargeVoltage of 19.200V. All numbers requesting charge voltage below 1.024V result in a voltage set point of zero, which terminates charging. Upon initial power-up or reset, the ChargeVoltage and ChargeCurrent registers are reset to 0 and the charger remains shut down until valid numbers are sent to the ChargeVoltage and ChargeCurrent registers. Use the Write-Word protocol (Figure 19) to write to the ChargeVoltage register. The register address for ChargeVoltage is 0x15. The 16-bit binary number formed by D15-D0 represents the charge voltage set point in mV. However, the resolution of the ISL88731A is 16mV because the D0-D3 bits are ignored as shown in Table 2. The D-5 bit is also ignored because it is not needed to span the 1.024V to 19.2V range. Table 2 shows the mapping between the charge-voltage set point and the 16-bit number written to the ChargeVoltage register. The ChargeVoltage register can be read back to verify its contents.
TABLE 2. CHARGEVOLTAGE (REGISTER 0x15) BIT
0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 Charge Voltage, DACV 0 Charge Voltage, DACV 1 Charge Voltage, DACV 2 Charge Voltage, DACV 3 Charge Voltage, DACV 4 Charge Voltage, DACV 5 Charge Voltage, DACV 6 Charge Voltage, DACV 7 Charge Voltage, DACV 8 Charge Voltage, DACV 9 Charge Voltage, DACV 10
BIT NAME
Not used Not used Not used Not used
DESCRIPTION
0 = Adds 0mV of charger voltage, 1024mV min 1 = Adds 16mV of charger voltage 0 = Adds 0mV of charger voltage, 1024mV min 1 = Adds 32mV of charger voltage 0 = Adds 0mV of charger voltage, 1024mV min 1 = Adds 64mV of charger voltage 0 = Adds 0mV of charger voltage, 1024mV min 1 = Adds 128mV of charger voltage 0 = Adds 0mV of charger voltage, 1024mV min 1 = Adds 256mV of charger voltage 0 = Adds 0mV of charger voltage, 1024mV min 1 = Adds 512mV of charger voltage 0 = Adds 0mA of charger voltage 1 = Adds 1024mV of charger voltage 0 = Adds 0mV of charger voltage 1 = Adds 2048mV of charger voltage 0 = Adds 0mV of charger voltage 1 = Adds 4096mV of charger voltage 0 = Adds 0mV of charger voltage 1 = Adds 8192mV of charger voltage 0 = Adds 0mV of charger voltage 1 = Adds 16384mV of charger voltage, 19200mV max Not used. Normally a 32768mV weight
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ISL88731A Setting Charge Current
ISL88731A has a 16-bit ChargeCurrent register that sets the battery charging current. ISL88731A controls the charge current by controlling the CSOP-CSON voltage. The register's LSB translates to 10V at CSON-CSOP. With a 10m charge current Rsense resistor (RS2 in Figure 2), the LSB translates to 1mA charge current. The ISL88731A ignores the first 7 LSBs and uses the next 6 bits to control the current DAC. The charge-current range of the ISL88731A is 0 to 8.064A (using a 10m current-sense resistor). All numbers requesting charge current above 8.064A result in a current setting of 8.064A. All numbers requesting charge current between 0mA to 128mA result in a current setting of 0mA. The default charge current setting at Power-On Reset (POR) is 0mA. To stop charging, set ChargeCurrent to 0. Upon initial power-up, the ChargeVoltage and ChargeCurrent registers are reset to 0 and the charger is disabled. To start the charger, write valid numbers to the ChargeVoltage and ChargeCurrent registers. The ChargeCurrent register uses the Write-Word protocol (see Figure 19). The register code for ChargeCurrent is 0x14 (0b00010100). Table 3 shows the mapping between the charge current set point and the ChargeCurrent number. The ChargeCurrent register can be read back to verify its contents. The ISL88731A includes a fault limiter for low battery conditions. If the battery voltage is less than 2.5V, the charge current is temporarily set to 128mA. The ChargeCurrent register is preserved and becomes active again when the battery voltage is higher than 2.7V. This function effectively provides a foldback current limit, which protects the charger during short circuit and overload.
TABLE 3. CHARGE CURRENT (REGISTER 0x14) (10m SENSE RESISTOR, RS2) BIT
0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 Charge Current, DACI 0 Charge Current, DACI 1 Charge Current, DACI 2 Charge Current, DACI 3 Charge Current, DACI 4 Charge Current, DACI 5
BIT NAME
Not used Not used Not used Not used Not used Not used Not used 0 = Adds 0mA of charger current 1 = Adds 128mA of charger current 0 = Adds 0mA of charger current 1 = Adds 256mA of charger current 0 = Adds 0mA of charger current 1 = Adds 512mA of charger current
DESCRIPTION
0 = Adds 0mA of charger current 1 = Adds 1024mA of charger current 0 = Adds 0mA of charger current 1 = Adds 2048mA of charger current 0 = Adds 0mA of charger current 1 = Adds 4096mA of charger current, 8064mA max Not used Not used Not used
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ISL88731A Setting Input-Current Limit
The total power from an AC-adapter is the sum of the power supplied to the system and the power into the charger and battery. When the input current exceeds the set input current limit, the ISL88731A decreases the charge current to provide priority to system load current. As the system load rises, the available charge current drops linearly to zero. Thereafter, the total input current can increase to the limit of the AC-adapter. The internal amplifier compares the differential voltage between CSSP and CSSN to a scaled voltage set by the InputCurrent register. The total input current is the sum of the device supply current, the charger input current, and the system load current. The total input current can be estimated by using Equation 2.
I INPUT = I SYSTEM + [ ( I CHARGE x V BATTERY ) ( V IN x ) ] (EQ. 2)
To set the input current limit use the SMBus to write a 16-bit InputCurrent register using the data format listed in Table 4. The InputCurrent register uses the Write-Word protocol (see Figure 19). The register code for InputCurrent is 0x3F (0b00111111). The InputCurrent register can be read back to verify its contents. The ISL88731A ignores the first 7 LSBs and uses the next 6 bits to control the input-current DAC. The input-current range of the ISL88731A is from 256mA to 11.004A. All 16-bit numbers requesting input current above 11.004A result in an input-current setting of 11.004A. All 16-bit numbers requesting input current between 0mA to 256mA result in an input-current setting of 0mA. The default input-current-limit setting at POR is 256mA. When choosing the current-sense resistor RS1, carefully calculate its power rating. Take into account variations in the system's load current and the overall accuracy of the sense amplifier. Note that the voltage drop across RS1 contributes additional power loss, which reduces efficiency. System currents normally fluctuate as portions of the system are powered up or put to sleep. Without input current regulation, the input source must be able to deliver the maximum system current and the maximum charger-input current. By using the input-current-limit circuit, the output-current capability of the AC wall adapter can be lowered, reducing system cost.
Where is the efficiency of the DC/DC converter (typically 85% to 95%). The ISL88731A has a 16-bit InputCurrent register that translates to a 2mA LSB and a 131.071A full-scale current using a 10m current-sense resistor (RS1 in Figure 2). Equivalently, the 16-bit InputCurrent number sets the voltage across CSSP and CSSN inputs in 20V per LSB increments.
TABLE 4. INPUT CURRENT (REGISTER 0x3F) (10m SENSE RESISTOR, RS1) BIT
0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 Input Current, DACS 0 Input Current, DACS 1 Input Current, DACS 2 Input Current, DACS 3 Input Current, DACS 4 Input Current, DACS 5
BIT NAME
Not used Not used Not used Not used Not used Not used Not used 0 = Adds 0mA of input current 1 = Adds 256mA of input current 0 = Adds 0mA of input current 1 = Adds 512mA of input current 0 = Adds 0mA of input current 1 = Adds 1024mA of input current 0 = Adds 0mA of input current 1 = Adds 2048mA of input current 0 = Adds 0mA of input current 1 = Adds 4096mA of input current
DESCRIPTION
0 = Adds 0mA of input current 1 = Adds 8192mA of input current, 11004mA max Not used Not used Not used
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FN6738.1 January 7, 2009
ISL88731A Charger Timeout
The ISL88731A includes 2 timers to insure the SMBus master is active and to prevent overcharging the battery. ISL88731A will terminate charging if the charger has not received a write to the ChargeVoltage or ChargeCurrent register within 175s or if the SCL line is low for more than 25ms. If a time-out occurs, either ChargeVoltage or ChargeCurrent registers must be written to re-enable charging. the lower byte stored in that register. After receiving the byte, the master Acknowledges by holding SDA low during the 9th clock pulse. ISL88731A then sends the higher byte stored in the register. After the second byte, neither device holds SDA low (No Acknowledge). The master will then produce a Stop condition to end the read transaction. ISL88731A does not support reading more than 1 register per transaction.
ISL88731A Data Byte Order
Each register in ISL88731A contains 16bits or 2, 8 bit bytes. All data sent on the SMBus is in 8-bit bytes and 2 bytes must be written or read from each register in ISL88731A. The order in which these bytes are transmitted appears reversed from the way they are normally written. The LOW byte is sent first and the HI byte is sent second. For example, When writing 0x41A0, 0xA0 is written first and 0x41 is sent second.
Application Information
The following battery charger design refers to the "Typical Application Circuit" (see Figure 2) on page 2, where typical battery configuration of 3S2P is used. This section describes how to select the external components including the inductor, input and output capacitors, switching MOSFETs and current sensing resistors.
Inductor Selection
The inductor selection has trade-offs between cost, size, crossover frequency and efficiency. For example, the lower the inductance, the smaller the size, but ripple current is higher. This also results in higher AC losses in the magnetic core and the windings, which decreases the system efficiency. On the other hand, the higher inductance results in lower ripple current and smaller output filter capacitors, but it has higher DCR (DC resistance of the inductor) loss, lower saturation current and has slower transient response. So, the practical inductor design is based on the inductor ripple current being 15% to 20% of the maximum operating DC current at maximum input voltage. Maximum ripple is at 50% duty cycle or VBAT = VIN,MAX/2. The required inductance for 15% ripple current can be calculated from Equation 3:
V IN, MAX L = -------------------------------------------------------4 F SW 0.3 I L, MAX (EQ. 3)
Writing to the Internal Registers
In order to set the charge current, charge voltage or input current, valid 16-bit numbers must be written to ISL88731A's internal registers via the SMBus. To write to a register in the ISL88731A, the master sends a control byte with the R/W bit set to 0, indicating a write. If it receives an Acknowledge from the ISL88731A it sends a register address byte setting the register to be written (i.e. 0x14 for the ChargeCurrent register). The ISL88731A will respond with an Acknowledge. The master then sends the lower data byte to be written into the desired register. The ISL88731A will respond with an Acknowledge. The master then sends the higher data byte to be written into the desired register. The ISL88731A will respond with an Acknowledge. The master then issues a Stop condition, indicating to the ISL88731A that the current transaction is complete. Once this transaction completes, the ISL88731A will begin operating at the new current or voltage. ISL88731A does not support writing more than one register per transaction.
Where, VIN,(MAX) is the maximum input voltage, FSW is the switching frequency and IL,(MAX) is the max DC current in the inductor. For VIN,(MAX) = 20V, VBAT = 12.6V, IBAT,(MAX) = 4.5A, and fs = 400kHz, the calculated inductance is 9.3H. Choosing the closest standard value gives L = 10H. Ferrite cores are often the best choice since they are optimized at 400kHz to 600kHz operation with low core loss. The core must be large enough not to saturate at the peak inductor current IPeak in Equation 4:
1 I PEAK = I L, MAX + -- I RIPPLE 2 (EQ. 4)
Reading from the Internal Registers
The ISL88731A has the ability to read from 5 internal registers. Prior to reading from an internal register, the master must first select the desired register by writing to it and sending the registers address byte. This process begins by the master sending a control byte with the R/W bit set to 0, indicating a write. Once it receives an Acknowledge from the ISL88731A it sends a register address byte representing the internal register it wants to read. The ISL88731A will respond with an Acknowledge. The master must then respond with a Stop condition. After the Stop condition, the master follows with a new Start condition, then sends a new control byte with the ISL88731A slave address and the R/W bit set to 1, indicating a read. The ISL88731A will Acknowledge then send
Inductor saturation can lead to cascade failures due to very high currents. Conservative design limits the peak and RMS current in the inductor to less than 90% of the rated saturation current. Crossover frequency is heavily dependent on the inductor value. FCO should be less than 20% of the switching
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FN6738.1 January 7, 2009
ISL88731A
frequency and a conservative design has FCO less than 10% of the switching frequency. The highest FCO is in voltage control mode with the battery removed and may be calculated (approximately) from Equation 5:
5 11 R SENSE F CO = -----------------------------------------2 L
(EQ. 5)
Output Capacitor Selection
The output capacitor in parallel with the battery is used to absorb the high frequency switching ripple current and smooth the output voltage. The RMS value of the output ripple current IRMS is given by Equation 6:
V IN, MAX I RMS = ----------------------------------- D ( 1 - D ) 12 L F SW
charger application, the input voltage of the synchronous buck converter is equal to the AC-adapter output voltage, which is relatively constant. The maximum efficiency is achieved by selecting a high-side MOSFET that has the conduction losses equal to the switching losses. Switching losses in the low-side FET are very small. The choice of low-side FET is a trade-off between conduction losses (rDS(ON)) and cost. A good rule of thumb for the rDS(ON) of the low-side FET is 2x the rDS(ON) of the high-side FET. The LGATE gate driver can drive sufficient gate current to switch most MOSFETs efficiently. However, some FETs may exhibit cross conduction (or shoot-through) due to current injected into the drain-to-source parasitic capacitor (Cgd) by the high dV/dt rising edge at the phase node when the high-side MOSFET turns on. Although LGATE sink current (1.8A typical) is more than enough to switch the FET off quickly, voltage drops across parasitic impedances between LGATE and the MOSFET can allow the gate to rise during the fast rising edge of voltage on the drain. MOSFETs with low threshold voltage (<1.5V) and low ratio of Cgs/Cgd (<5) and high gate resistance (>4) may be turned on for a few ns by the high dV/dt (rising edge) on their drain. This can be avoided with higher threshold voltage and Cgs/Cgd ratio. Another way to avoid cross conduction is slowing the turn-on speed of the high-side MOSFET by connecting a resistor between the BOOT pin and the boot strap capacitor. For the high-side MOSFET, the worst-case conduction losses occur at the minimum input voltage, as shown in Equation 8:
V OUT 2 P Q1, conduction = --------------- I BAT r DS ( ON ) V IN (EQ. 8)
(EQ. 6)
Where the duty cycle D is the ratio of the output voltage (battery voltage) over the input voltage for continuous conduction mode which is typical operation for the battery charger. During the battery charge period, the output voltage varies from its initial battery voltage to the rated battery voltage. So, the duty cycle varies from 0.53 for the minimum battery voltage of 7.5V (2.5V/Cell) to 0.88 for the maximum battery voltage of 12.6V. The maximum RMS value of the output ripple current occurs at the duty cycle of 0.5 and is expressed as Equation 7:
V IN, MAX I RMS = ------------------------------------------4 12 L F SW
(EQ. 7)
For VIN,(MAX) = 19V, VBAT = 16.8V, L = 10H, and fs = 400kHz, the maximum RMS current is 0.19A. A typical 20F ceramic capacitor is a good choice to absorb this current and also has very small size. Organic polymer capacitors have high capacitance with small size and have a significant equivalent series resistance (ESR). Although ESR adds to ripple voltage, it also creates a high frequency zero that helps the closed loop operation of the buck regulator. EMI considerations usually make it desirable to minimize ripple current in the battery leads. Beads may be added in series with the battery pack to increase the battery impedance at 400kHz switching frequency. Switching ripple current splits between the battery and the output capacitor depending on the ESR of the output capacitor and battery impedance. If the ESR of the output capacitor is 10m and battery impedance is raised to 2 with a bead, then only 0.5% of the ripple current will flow in the battery.
The optimum efficiency occurs when the switching losses equal the conduction losses. However, it is difficult to calculate the switching losses in the high-side MOSFET since it must allow for difficult-to-quantify factors that influence the turn-on and turn-off times. These factors include the MOSFET internal gate resistance, gate charge, threshold voltage, stray inductance and the pull-up and pulldown resistance of the gate driver. The following switching loss calculation (Equation 9) provides a rough estimate.
P Q1, Switching = Q gd 1 Q gd 1 -- V IN I LV f sw ------------------------ + -- V IN I LP f sw ---------------- + Q rr V IN f sw I g, source 2 2 I g, sin k
(EQ. 9)
MOSFET Selection
The Notebook battery charger synchronous buck converter has the input voltage from the AC-adapter output. The maximum AC-adapter output voltage does not exceed 25V. Therefore, 30V logic MOSFET should be used. The high-side MOSFET must be able to dissipate the conduction losses plus the switching losses. For the battery
Where, the following are the peak gate-drive source/sink current of Q1, respectively: * Qgd: drain-to-gate charge, * Qrr: total reverse recovery charge of the body-diode in low-side MOSFET, * ILV: inductor valley current, * ILP: Inductor peak current, * Ig,sink * Ig,source
FN6738.1 January 7, 2009
17
ISL88731A
Low switching loss requires low drain-to-gate charge Qgd. Generally, the lower the drain-to-gate charge, the higher the ON-resistance. Therefore, there is a trade-off between the ON-resistance and drain-to-gate charge. Good MOSFET selection is based on the Figure of Merit (FOM), which is a product of the total gate charge and ON-resistance. Usually, the smaller the value of FOM, the higher the efficiency for the same application. For the low-side MOSFET, the worst-case power dissipation occurs at minimum battery voltage and maximum input voltage (Equation 10):
V OUT 2 P Q2 = 1 - --------------- I BAT r DS ( ON ) V IN
Input Capacitor Selection
The input capacitor absorbs the ripple current from the synchronous buck converter, which is given by Equation 14:
Irms = IBAT VOUT (VIN - VOUT ) VIN
(EQ. 14)
(EQ. 10)
Choose a low-side MOSFET that has the lowest possible ON-resistance with a moderate-sized package like the SO-8 and is reasonably priced. The switching losses are not an issue for the low-side MOSFET because it operates at zero-voltage-switching. Ensure that the required total gate drive current for the selected MOSFETs should be less than 24mA. So, the total gate charge for the high-side and low-side MOSFETs is limited by Equation 11:
I GATE Q GATE ---------------f sw
This RMS ripple current must be smaller than the rated RMS current in the capacitor datasheet. Non-tantalum chemistries (ceramic, aluminum, or OSCON) are preferred due to their resistance to power-up surge currents when the AC-adapter is plugged into the battery charger. For Notebook battery charger applications, it is recommended that ceramic capacitors or polymer capacitors from Sanyo be used due to their small size and reasonable cost.
Loop Compensation Design
ISL88731A has three closed loop control modes. One controls the output voltage when the battery is fully charged or absent. A second controls the current into the battery when charging and the third limits current drawn from the adapter. The charge current and input current control loops are compensated by a single capacitor on the ICOMP pin. The voltage control loop is compensated by a network on the VCOMP pin. Descriptions of these control loops and guidelines for selecting compensation components will be given in the following sections. Which loop controls the output is determined by the minimum current buffer and the minimum voltage buffer shown in the "Functional Block Diagram" (see Figure 1) on page 2. These three loops will be described separately.
(EQ. 11)
Where IGATE is the total gate drive current and should be less than 24mA. Substituting IGATE = 24mA and fs = 400kHz into Equation 11 yields that the total gate charge should be less than 80nC. Therefore, the ISL88731A easily drives the battery charge current up to 8A.
Transconductance Amplifiers GMV, GMI and GMS
ISL88731A uses several transconductance amplifiers (also known as gm amps). Most commercially available op amps are voltage controlled voltage sources with gain expressed as A = VOUT/VIN. gm amps are voltage controlled current sources with gain expressed as gm = IOUT/VIN. gm will appear in some of the equations for poles and zeros in the compensation.
Snubber Design
ISL88731A's buck regulator operates in discontinuous current mode (DCM) when the load current is less than half the peak-to-peak current in the inductor. After the low-side FET turns off, the phase voltage rings due to the high impedance with both FETs off. This can be seen in Figure 9. Adding a snubber (resistor in series with a capacitor) from the phase node to ground can greatly reduce the ringing. In some situations a snubber can improve output ripple and regulation. The snubber capacitor should be approximately twice the parasitic capacitance on the phase node. This can be estimated by operating at very low load current (100mA) and measuring the ringing frequency. CSNUB and RSNUB can be calculated from Equations 12 and 13:
2 C SNUB = -----------------------------------2 ( 2F ring ) L (EQ. 12) R SNUB = 2L ------------------C SNUB (EQ. 13)
PWM Gain Fm
The Pulse Width Modulator in the ISL88731A converts voltage at VCOMP to a duty cycle by comparing VCOMP to a triangle wave (duty = VCOMP/VP-P RAMP). The low-pass filter formed by L and CO convert the duty cycle to a DC output voltage (Vo = VDCIN*duty). In ISL88731A, the triangle wave amplitude is proportional to VDCIN. Making the ramp amplitude proportional to DCIN makes the gain from VCOMP to the PHASE output a constant 11 and is independent of DCIN. For small signal AC analysis, the battery is modeled by its internal resistance. The total output resistance is the sum of the sense resistor and the internal resistance of the MOSFETs, inductor and capacitor. Figure20 shows the small signal model of the pulse width modulator (PWM), power stage, output filter and battery.
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ISL88731A
PHASE PHASE 11 11
VDD RAMP GEN VRAMP = VDD/11
L RFET_RDSON RFET_RDSON RL_DCR RL_DCR
+
PWM INPUT
DRIVERS
L
CO
S S
0.25
+
CA2 20X
CSOP CSOP +
RF2 RF2 CF2 CF2 RS2 RS2
-
CSON CSON
ICOMP ICOMP CICOMP CICOMP
GMI GMI
+ +
DACI DACI
CO CO RESR RESR
RBAT RBAT
PWM GAIN=11 11 RFET_RDSON PWM INPUT
L
RL_DCR
RSENSE
FIGURE 21. CHARGE CURRENT LIMIT LOOP
CO RBAT RESR
frequency (smaller CICOMP), the DC gain will be higher but the phase margin will be lower. Use a capacitor on ICOMP that is equal to or greater than the value calculated in Equation 18. The factor of 1.5 is to ensure the zero is at a frequency lower than the pole including tolerance variations.
1.5 4 ( 50A V ) L C ICOMP = -----------------------------------------------------------------------------------------------------( R SENSE + r DS ( ON ) + R DCR + R BAT ) (EQ. 18)
FIGURE 20. SMALL SIGNAL AC MODEL
In most cases the Battery resistance is very small (<200m) resulting in a very low Q in the output filter. This results in a frequency response from the input of the PWM to the inductor current with a single pole at the frequency calculated in Equation 15:
( R SENSE + r DS ( ON ) + R DCR + R BAT ) F POLE1 = -----------------------------------------------------------------------------------------------------2 L (EQ. 15)
A filter should be added between RS2 and CSOP and CSON to reduce switching noise. The filter roll-off frequency should be between the crossover frequency and the switching frequency (~100kHz). RF2 should be small (<10) to minimize offsets due to leakage current into CSOP. The filter cut off frequency is calculated using Equation 19:
1 F FILTER = -----------------------------------------( 2 C F2 R F2 )
The output capacitor creates a pole at a very high frequency due to the small resistance in parallel with it. The frequency of this pole is calculated in Equation 16:
1 F POLE2 = -------------------------------------2 C o R BAT
(EQ. 19)
(EQ. 16)
Charge Current Control Loop When the battery is less than the fully charged, the voltage error amplifier goes to it's maximum output (limited to 0.3V above ICOMP) and the ICOMP voltage controls the loop through the minimum voltage buffer. Figure 22 shows the charge current control loop. The compensation capacitor (CICOMP) gives the error amplifier (GMI) a pole at a very low frequency (<<1Hz) and a a zero at FZ1. FZ1 is created by the 0.25*CA2 output added to ICOMP. The frequency can be calculated from Equation 17:
4 gm2 F ZERO = --------------------------------------( 2 C ICOMP ) gm2 = 50A V (EQ. 17)
The crossover frequency is determined by the DC gain of the modulator and output filter and the pole in Equation 16. The DC gain is calculated in Equation 20 and the crossover frequency is calculated with Equation 21:
11 R SENSE A DC = -----------------------------------------------------------------------------------------------------( R SENSE + r DS ( ON ) + R DCR + R BAT ) 11 R SENSE F CO = A DC F POLE = ---------------------------------2 L
(EQ. 20)
(EQ. 21)
The Bode plot of the loop gain, the compensator gain and the power stage gain is shown in Figure 22.
Adapter Current Limit Control Loop
If the combined battery charge current and system load current draws current that equals the adapter current limit set by the InputCurrent register, ISL88731A will reduce the current to the battery and/or reduce the output voltage to hold the adapter current at the limit. Above the adapter current limit, the minimum current buffer equals the output of GMS and ICOMP controls the charger output. Figure 23 shows the adapter current limit control loop.
Placing this zero at a frequency equal to the pole calculated in Equation 16 will result in maximum gain at low frequencies and phase margin near 90. If the zero is at a higher
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ISL88731A
ChargeVoltage register the voltage error amplifier (GMV) takes control of the output (assuming that the adapter current is below the limit set by ACLIM). The voltage error amplifier (GMV) discharges the capacitor on VCOMP to limit the output voltage. The current to the battery decreases as the cells charge to the fixed voltage and the voltage across the internal battery resistance decreases. As battery current decreases the two current error amplifiers (GMI and GMS) output their maximum current and charge the capacitor on ICOMP to its maximum voltage (limited to 0.3V above VCOMP). With high voltage on ICOMP, the minimum voltage buffer output equals the voltage on VCOMP. The voltage control loop is shown in Figure 24.
F
-60 0.01 0.1 1 10
POLE2
60 Compensator Modulator Loop
40
F
ZERO
20 GAIN (dB)
0
-20
F
POLE1
F
FILTER
-40
PHASE
100 1000
L
11 RFET_RDSON RL_DCR
FREQUENCY (kHz)
FIGURE 22. CHARGE CURRENT LOOP BODE PLOTS
DCIN
RS1 11 RF1 CF1 RFET_RDSON RL_DCR PHASE L
VCOMP
S
+ 0.25 -
CA2
CSOP
RF2 CF2 RS2
+
20x
CSON
GMV
-
R3 CO R4 RESR RBAT
+
S
CSSN
+
0.25
CA2
CSOP +
RF2 CF2 RS2
CVCOMP DACV RVCOMP
-
20X
CSON
- 20
+
CA1
CO RESR
CSSP
RBAT
FIGURE 24. VOLTAGE CONTROL LOOP
GMS
ICOMP CICOMP + DACS
Output LC Filter Transfer Functions
The gain from the phase node to the system output and battery depend entirely on external components. Typical output LC filter response is shown in Figure 25. Transfer function ALC(s) is shown in Equation 22:
s 1 - --------------- ESR A LC = --------------------------------------------------------- s2 s ----------- + ------------------------ + 1 DP ( LC Q ) 1 ESR = -------------------------------( R ESR C o ) 1 LC = ----------------------( L Co ) L Q = R o -----Co (EQ. 22)
FIGURE 23. ADAPTER CURRENT LIMIT CONTROL LOOP
The loop response equations, bode plots and the selection of CICOMP are the same as the charge current control loop with loop gain reduced by the duty cycle and the ratio of RS1/RS2. In other words, if RS1 = RS2 and the duty cycle D = 50%, the loop gain will be 6dB lower than the loop gain in Figure 23. This gives lower crossover frequency and higher phase margin in this mode. If RS1/RS2 = 2 and the duty cycle is 50% then the adapter current loop gain will be identical to the gain in Figure 23. A filter should be added between RS1 and CSIP and CSIN to reduce switching noise. The filter roll off frequency should be between the crossover frequency and the switching frequency (~100kHz).
The resistance RO is a combination of MOSFET rDS(ON), inductor DCR, RSENSE and the internal resistance of the battery (normally between 50m and 200m) The worst case for voltage mode control is when the battery is absent. This results in the highest Q of the LC filter and the lowest phase margin. The compensation network consists of the voltage error amplifier GMV and the compensation network RVCOMP,
Voltage Control Loop
When the battery is charged to the voltage set by
20
FN6738.1 January 7, 2009
ISL88731A
CVCOMP which give the loop very high DC gain, a very low frequency pole and a zero at FZERO1. Inductor current information is added to the feedback to create a second zero FZERO2. The low pass filter RF2, CF2 between RS2 and ISL88731A add a pole at FFILTER. R3 and R4 are internal divider resistors that set the DC output voltage. For a 3-cell battery, R3 = 500k and R4 = 100k. Equations 23 through 30 relate the compensation network's poles, zeros and gain to the components in Figure 24. Figure 26 shows an asymptotic Bode plot of the DC/DC converter's gain vs frequency. It is strongly recommended that FZERO1 is approximately 30% of FLC and FZERO2 is approximately 70% of FLC.
Compensation Break Frequency Equations
1 F ZERO1 = ---------------------------------------------------------------------( 2 C VCOMP R 1COMP ) R VCOMP R 4 gm1 F ZERO2 = ---------------------------------------------- -------------------- ----------- 2 R SENSE C o R 4 + R 3 5 (EQ. 24) 1 F LC = ------------------------------( 2 L C o ) 1 F FILTER = -----------------------------------------( 2 R F2 C F2 ) 1 F POLE1 = --------------------------------------------------( 2 R SENSE C o ) (EQ. 25)
(EQ. 23)
(EQ. 26) (EQ. 27)
NO BATTERY
GAIN (dB)
1 F ESR = ------------------------------------------( 2 C o R ESR )
(EQ. 28)
RBATTERY = 200m RBATTERY = 50m
Choose RVCOMP equal or lower than the value calculated from Equation 29.
R 3 + R 4 5 R VCOMP = ( 0.7 F LC ) ( 2 C o R SENSE ) ----------- -------------------- gm1 R 4 (EQ. 29)
Next choose CVCOMP equal or higher than the value calculated from Equation 30.
PHASE ()
1 C VCOMP = -------------------------------------------------------------------------( 0.3 F LC ) ( 2 R VCOMP )
PCB Layout Considerations
(EQ. 30)
Power and Signal Layers Placement on the PCB
FREQUENCY
FIGURE 25. FREQUENCY RESPONSE OF THE LC OUTPUT FILTER
As a general rule, power layers should be close together, either on the top or bottom of the board, with signal layers on the opposite side of the board. As an example, layer arrangement on a 4-layer board is shown below: 1. Top Layer: signal lines, or half board for signal lines and the other half board for power lines
60 Compensator 40 Modulator Loop
2. Signal Ground
F FPOLE1
3. Power Layers: Power Ground 4. Bottom Layer: Power MOSFET, Inductors and other Power traces Separate the power voltage and current flowing path from the control and logic level signal path. The controller IC will stay on the signal layer, which is isolated by the signal ground to the power signal traces.
20 GAIN (dB)
LC
0
FFILTER
-20
F
-40
ZERO1
Component Placement
F
ZERO2
F
ESR
-60 0.1 1 10 FREQUENCY (kHz) 100 1000
The power MOSFET should be close to the IC so that the gate drive signal, the LGATE, UGATE, PHASE, and BOOT, traces can be short. Place the components in such a way that the area under the IC has less noise traces with high dv/dt and di/dt, such as gate signals and phase node signals.
FIGURE 26. ASYMPTOTIC BODE PLOT OF THE VOLTAGE CONTROL LOOP GAIN
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FN6738.1 January 7, 2009
ISL88731A
Signal Ground and Power Ground Connection
At minimum, a reasonably large area of copper, which will shield other noise couplings through the IC, should be used as signal ground beneath the IC. The best tie-point between the signal ground and the power ground is at the negative side of the output capacitor on each side, where there is little noise; a noisy trace beneath the IC is not recommended.
HIGH CURRENT TRACE SENSE RESISTOR HIGH CURRENT TRACE
GND and VCC Pin
At least one high quality ceramic decoupling capacitor should be used to cross these two pins. The decoupling capacitor can be put close to the IC.
KELVIN CONNECTION TRACES TO THE LOW PASS FILTER AND CSOP AND CSON
FIGURE 27. CURRENT SENSE RESISTOR LAYOUT
LGATE Pin
This is the gate drive signal for the bottom MOSFET of the buck converter. The signal going through this trace has both high dv/dt and high di/dt, and the peak charging and discharging current is very high. These two traces should be short, wide, and away from other traces. There should be no other traces in parallel with these traces on any layer.
CSOP, CSON, CSSP and CSSN Pins
Accurate charge current and adapter current sensing is critical for good performance. The current sense resistor connects to the CSON and the CSOP pins through a low pass filter with the filter capacitor very near the IC (see Figure 2). Traces from the sense resister should start at the pads of the sense resister and should be routed close together, through the low pass filter and to the CSOP and CSON pins (see Figure 27). The CSON pin is also used as the battery voltage feedback. The traces should be routed away from the high dv/dt and di/dt pins like PHASE, BOOT pins. In general, the current sense resistor should be close to the IC. These guidelines should also be followed for the adapter current sense resister and CSSP and CSSN. Other layout arrangements should be adjusted accordingly.
PGND Pin
PGND pin should be laid out to the negative side of the relevant output capacitor with separate traces.The negative side of the output capacitor must be close to the source node of the bottom MOSFET. This trace is the return path of LGATE.
PHASE Pin
This trace should be short, and positioned away from other weak signal traces. This node has a very high dv/dt with a voltage swing from the input voltage to ground. No trace should be in parallel with it. This trace is also the return path for UGATE. Connect this pin to the high-side MOSFET source.
DCIN Pin
This pin connects to AC-adapter output voltage, and should be less noise sensitive.
Copper Size for the Phase Node
The capacitance of PHASE should be kept very low to minimize ringing. It would be best to limit the size of the PHASE node copper in strict accordance with the current and thermal management of the application.
UGATE Pin
This pin has a square shape waveform with high dv/dt. It provides the gate drive current to charge and discharge the top MOSFET with high di/dt. This trace should be wide, short, and away from other traces, similar to the LGATE.
Identify the Power and Signal Ground
The input and output capacitors of the converters, the source terminal of the bottom switching MOSFET PGND should connect to the power ground. The other components should connect to signal ground. Signal and power ground are tied together at one point.
BOOT Pin
This pin's di/dt is as high as the UGATE; therefore, this trace should be as short as possible.
Clamping Capacitor for Switching MOSFET
It is recommended that ceramic capacitors be used closely connected to the drain of the high-side MOSFET, and the
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For information regarding Intersil Corporation and its products, see www.intersil.com 22
FN6738.1 January 7, 2009
ISL88731A
Package Outline Drawing
L28.5x5B
28 LEAD THIN QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 1, 10/07
4X 3.0 5.00 A B 6 PIN 1 INDEX AREA 24X 0.50 6 PIN #1 INDEX AREA
22
28 1
21
5.00
3 .25 0 . 10
15
(4X) 0.15 14 8
7
TOP VIEW
28X 0.55 0.05
0.10 M C A B 4 28X 0.25 0.05
BOTTOM VIEW
SEE DETAIL "X"
0.10 C
0 . 75 0.05 BASE PLANE ( 4. 65 TYP ) ( 24X 0 . 50) ( 3. 25)
C
SEATING PLANE 0.08 C
SIDE VIEW
(28X 0 . 25 ) C 0 . 2 REF 5
( 28X 0 . 75)
0 . 00 MIN. 0 . 05 MAX.
TYPICAL RECOMMENDED LAND PATTERN
DETAIL "X"
NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal 0.05 4. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 identifier may be either a mold or mark feature.
23
FN6738.1 January 7, 2009

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